Biasing and driving circuit, based on a feedback voltage regulator, for an electric load

ABSTRACT

A biasing and driving circuit for an electric load, having an input adapted to receive an a.c. input voltage and an output adapted to supply a d.c. output voltage, comprising: a voltage-regulator device, having a feedback input terminal configured to receive a sensing voltage that is a function of a supply current that flows through the electric load and regulating, on the basis of the sensing voltage received, the supply current; a resistive sensing element, operatively coupled to the feedback input, configured to receive the supply current and generate the sensing voltage as a function of the supply current; a resistor coupled to the feedback input; and an auxiliary biasing circuit adapted to receive the a.c. input voltage and inject through the resistor an a.c. auxiliary biasing current that varies in a way inversely proportional to the input voltage.

BACKGROUND

1. Technical Field

The present disclosure relates to a biasing and driving circuit, basedon a feedback voltage regulator, in particular a switching-mode powersupply (SMPS), for an electric load.

2. Description of the Related Art

FIG. 1A shows a driving circuit for light-emitting diodes (LEDs), whichis designated by the reference 1 and in particular includes a voltageregulator 5 of an SMPS type configured to operate as constant-currentsource and adapted to supply a string of LEDs 2 (illustrated in FIG. 1Ais just one LED 2, coupled between output supply pins, V_(LED) ⁺ andV_(LED) ⁻). The SMPS device 5 is of a per se known type, for exampledescribed in the datasheet of the product “LED5000” manufactured bySTMicroelectronics, entitled “LED5000—A monolithic step-down currentsource with dimming capability”, September, 2014.

The SMPS device 5 is operatively coupled to input supply terminalsV_(IN) ⁺ and V_(IN) ⁻, present between which is a voltage V_(IN), forexample generated by an electronic transformer (not illustrated).

The SMPS device 5 has, in a known way, a plurality of operatingterminals, and in particular: a supply terminal 1 a, adapted to receivean input voltage V_(IN), having a value, for example between 5.5 and 48V; a reference terminal 1 b, which forms a reference-voltage terminal; afeedback input terminal 1 c, which is coupled to the sensing resistor 4and constitutes the inverting input of an error amplifier internal tothe SMPS device 5 (regulation terminal); a terminal 1 d, which providesa power-supply connection for the internal analog circuitry; a terminal1 e, which forms, together with the reference terminal 1 b and with theerror amplifier, the output of a regulation loop internal to the SMPSdevice 5; and a terminal 1 f that implements an output terminal forswitching the SMPS device 5 and is coupled to the terminal 1 d via acapacitor 8.

As illustrated in greater detail in FIG. 1B, the regulation loopinternal to the SMPS device 5 includes the voltage-error amplifier 3,which implements a first stage of the regulation loop. In particular,the voltage-error amplifier 3 is a transconductance operationalamplifier, the non-inverting input of which is connected to a voltagereference V_(REF) internal to the SMPS device 5 (variation of which istypically between 194 and 206 mV; in particular, a typical value of 200mV in closed loop is considered in what follows), whereas the invertinginput terminal is connected to a sensing resistor 4. The inverting inputterminal of the voltage-error amplifier 3 forms a feedback inputterminal 1 c of the SMPS device 5. The voltage-error amplifier 3generates, on the terminal 1 c, a control signal V_(CONTROL), which issupplied to the non-inverting input of a PWM comparator 7, which, inturn, drives, on the terminal 1 f, the high-side (HS) switch of a DC-DCconverter 9′. A current detector 9″ detects the current circulating inthe high-side (HS) switch and supplies the value detected (transduced)to the inverting input of the PWM comparator.

The DC-DC converter 9′ generates at output a regulation signal V_(SW)having a duty cycle such as to regulate the supply current (I_(LED))appropriately.

In other words, present between the terminal 1 a and the terminal 1 f isan SMPS converter, wherein the non-inverting input of the erroramplifier acts on the terminal 1 c, and the output of the amplifier actson the terminal 1 e.

Thus, with reference to FIG. 1A, the SMPS regulator 5, the inductor 6and the diode 11 form, for example, a DC-DC converter topology of aboost type.

The regulated current level supplied at output from the SMPS device 5 isthus set, or regulated, on the basis of the current that flows throughthe sensing resistor 4, across which, according to what has been said,there may be noted a voltage drop equal to the reference V_(REF) of 200mV. The resistance value R_(S) of the sensing resistor 4 is consequentlygiven by R_(S)=(200 mV)/I_(LED), where I_(LED) is the current that flowsthrough the string of LEDs 2. In a case provided by way of example,where L_(LED)=1A, we have R_(S)=0.2 Ω.

Coupled to the terminal 1 c of the SMPS device 5 a resistor 26 isfurther present, having a resistance R₁ of approximately 10 kΩ.Optionally, it is possible to insert a Zener diode (not illustrated) inparallel to the resistor 26 so that the resistor 26 and the Zener diodeimplement a protection from overvoltages. The effect of the resistor 26is negligible in so far as the current at input to the terminal 1 c issubstantially zero, or negligible (at the most a few tens of nanoamps).

Furthermore, coupled to the terminal 1 e are a resistor 13 and acapacitor 15, connected together in series, which have the function ofimplementing a compensation network for the regulation loop. By way ofexample, the resistor 13 has a resistance of 22 kΩ and the capacitor 15has a capacitance of 10 nF.

It is evident that the SMPS device 5 may include further input/outputterminals, for implementing further functions, as required.

The input capacitor 10, coupled to respective supply inputs of the SMPSdevice 5, is configured to withstand the maximum operating input voltageand the maximum mean square value of the current. Capacitors adapted forthis purpose, available for use for a wide range of currents, are, forexample, electrolytic capacitors, ceramic capacitors, tantalumcapacitors.

An output capacitor 12, coupled between the input V_(IN) ⁺ and thereference terminal 1 b, has the function of filtering the current rippleof the diode 11, which, given a specific application and an outputcurrent, depends upon the value of inductance of the inductor 6. Ingeneral, if ΔI_(L) is the current ripple of the inductor 6 and I_(L) theaverage current that flows through the inductor, the value of inductanceL is chosen in such a way that (ΔI_(L)/I_(L))<0.5.

The driving circuit 1 may be coupled, as has been said, to an electronictransformer, which generates the input voltage V_(IN). Electronictransformers of a known type are typically based on a self-oscillatingcircuit and, to operate properly, require a load of a resistive type. Inother words, the driving circuit 1 must be seen, by an electronictransformer coupled to the inputs V_(IN) ⁺ and V_(IN) ⁻, as a resistiveload. However, it is known that an SMPS device, for example of the typeillustrated in FIG. 1A and described with reference to that figure, inthe absence of further arrangements, is seen as a load with negativeimpedance and thus is not optimized to be coupled to the output of anelectronic transformer that requires a resistive load for its properoperation.

To overcome this drawback, it is known in the art to use a currentcontrol of the input signal. See, for example, Application Note 5372,“MR16 LED Driver Makes MR16 LED Lamps Compatible with Most ElectronicTransformers” by Suresh Hariharan, Mar. 27, 2013, Maxim IntegratedProducts. A similar solution is discussed in the datasheet of theproduct MAX16840, manufactured by Maxim Integrated Products, Inc., “LEDDriver with Integrated MOSFET for MR16 and Other 12V AC Input Lamps”.

In this technical solution, represented schematically in FIG. 2, thevoltage on the sensing resistor 4 is regulated at each switching cycle,exploiting a reference circuit 18 external to the SMPS device 5, adaptedto supply a voltage signal V_(REFI) to a further input terminal 1 g ofthe SMPS device 5, for the purpose of setting the input current level byappropriately controlling the voltage on the terminal 1 c. In otherwords, when the voltage V_(REFI) present on the terminal 1 g drops belowa certain threshold value, the input current (voltage on the resistor 4)is regulated proportionally to the value assumed by the voltage V_(REFI)on the terminal 1 g. Instead, when the voltage V_(REFI) present on theterminal 1 g exceeds the threshold value, then the input current(voltage on the resistor 4) is set at a predefined fixed value. Thevoltage on the sensing resistor 4 is thus regulated as a function of thevoltage V_(REFI) received at input on the terminal 1 g, which is in turna function of the input voltage V_(IN). This type of modulation of thevoltage on the terminal 1 c enables simulation of a resistive load, seenby an electronic transformer coupled to the input of the driving circuit1 of FIG. 2. However, this implementation requires a terminal of thedevice 5 (terminal 1 g) explicitly dedicated to this purpose, acircuitry internal to the device 5 adapted for regulating the voltage onthe terminal 1 c as a function of the reference on the terminal 1 g, aswell as, at the same time, an external circuit for generating thereference signal to be supplied to the terminal 1 g. In other words,this solution is not applicable to any generic SMPS device; the latter,instead, must be purposely built.

Other known solutions require provision of dedicated dual-stageconverters, with consequent implementation of double inductivecomponents, which increase the costs and size.

There is thus a need to provide a driving circuit for a voltageregulator, for example of an SMPS type, that is adapted to emulate aresistive load when seen from the input terminals V_(IN) ⁺ and V_(IN) ⁻,is such as to increase the power factor, with lower production costs andreduced occupation of space, and is able to operate with any genericvoltage regulator.

BRIEF SUMMARY

An aim of the present disclosure is to provide a biasing and drivingcircuit, based on a voltage-regulator device, for an electric load,which overcomes problems of the known art and may achieve theaforementioned advantages.

Provided according to the present disclosure is a biasing and drivingcircuit, based on an SMPS device, of an electric load.

In one embodiment, a biasing and driving circuit for an electric loadhas input terminals adapted to receive an a.c. input voltage and outputterminals adapted to supply a d.c. output voltage to the electric load.A voltage regulator with a feedback loop has a feedback input terminalconfigured to receive a sensing voltage that is a function of a supplycurrent that flows through the electric load and which regulates, on thebasis of the sensing voltage, the supply current. A resistive sensingelement is operatively coupled to the feedback input and configured toreceive the supply current and generate the sensing voltage as afunction of the supply current. A current transducer, of a resistivetype, is coupled to the feedback input. An auxiliary biasing circuit isadapted to receive the a.c. input voltage and to inject through thecurrent transducer an a.c. auxiliary biasing current that varies in away inversely proportional to the a.c. input voltage.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

For a better understanding of the present disclosure, some embodimentsthereof will now be described, purely by way of non-limiting example andwith reference to the annexed drawings, wherein:

FIG. 1A illustrates a biasing and driving circuit for a string of LEDs,according to an embodiment of a known type;

FIG. 1B illustrates a regulation loop internal to an SMPS device thatforms part of the biasing and driving circuit of FIG. 1A;

FIG. 2 illustrates a biasing and driving circuit for a string of LEDs,according to a further embodiment of a known type;

FIG. 3 illustrates a biasing and driving circuit for a string of LEDs,according to one embodiment of the present disclosure;

FIG. 3a illustrates the circuit of FIG. 3 with an additional biasingcircuit that improves the behavior of the circuit of FIG. 3 according toanother embodiment of the present disclosure;

FIG. 4 illustrates a circuit implementation of the biasing and drivingcircuit of FIG. 3, according to an embodiment of the present disclosure;

FIG. 4a illustrates a circuit implementation of the biasing and drivingcircuit of FIG. 3a , according to an embodiment of the presentdisclosure;

FIGS. 5A-5H show electrical signals during operating steps of thebiasing and driving circuits of FIGS. 4 and 4 a;

FIG. 6 illustrates a further circuit implementation of the biasing anddriving circuit of FIG. 3, according to a further embodiment of thepresent disclosure;

FIG. 6a illustrates a further circuit implementation of the biasing anddriving circuit of FIG. 3a , according to a further embodiment of thepresent disclosure;

FIG. 6b illustrates a still further circuit implementation of thebiasing and driving circuit of FIG. 3a , according to yet anotherembodiment of the present disclosure; and

FIG. 7 illustrates, in greater detail, the biasing and driving circuitof FIG. 6.

DETAILED DESCRIPTION

FIG. 3 shows, according to an embodiment of the present disclosure, abiasing and driving circuit 20 for a string of LEDs, comprising aswitching-mode power-supply (SMPS) device 5, configured to operate asconstant-current source and adapted to supply the string of LEDs 2(illustrated by way of example in FIG. 3 is just one LED 2). Elements ofthe driving circuit 20 that are common to those of the driving circuit 1of FIG. 1A are designated by the same reference numbers and are notdescribed any further.

The driving circuit 20 further includes a current generator 22,operatively coupled to the terminal 1 c of the SMPS device 5, configuredto supply on the terminal 1 c of the SMPS device 5 a current signal I₁of an alternating current (a.c.) type, in particular a sinusoidalsignal. According to one aspect of the present disclosure, the bandwidthof the regulation loop internal to the SMPS device 5, illustrated inFIG. 1B, is greater than the maximum frequency of the current signal I₁,for example greater by one or more orders of magnitude. For instance,the bandwidth of the regulation loop of FIG. 1B is 10 kHz, and thefrequency of the current signal I₁ (e.g., a sinusoidal signal) is 100Hz.

The resistor 26, having a resistance R₁ with a value of approximately 10kΩ, is adapted to receive the current signal I₁, modulating the voltagedrop on the sensing resistor 4. The output terminal of the currentgenerator I₁ is coupled between the terminal 1 c and the resistor 26,and the reference terminal of the current generator, instead, is coupledto the input terminal V_(IN) ⁺. The terminal 1 c is a high-impedanceterminal, and consequently (to a first approximation) the current signalI₁ flows entirely through the resistor 26 and not towards the terminal 1c.

The current I_(LED) that flows through the sensing resistor 4 is, thus,given by:

$I_{LED} = \frac{V_{FB} - \left( {I_{1} \cdot R_{1}} \right)}{R_{S}}$where V_(FB) is the feedback voltage present on the terminal 1 c inclosed loop (in the example considered previously, equal to 200 mV) andI₁·R₁ is the voltage contribution generated by the resistor 26 in thepresence of the current signal I₁ supplied by the generator 22 (R₁ ishere chosen by way of example equal to 10 kΩ). In other words,V_(FB)−(I₁·R₁) is the voltage across the sensing resistor 4.

From the foregoing equation, it is evident that, in the absence of thecurrent signal I₁ (i.e., I₁=0 A), the current I_(LED)=V_(FB)/R_(S)circulating in the string of LEDs 2 and in the sensing resistor 4 isdetermined only by the internal reference V_(REF) (reference of theerror amplifier on the feedback terminal 1 c of the SMPS regulator 5).Instead, in the presence of the current signal I₁, the currentcirculating in the LEDs 2 depends upon the voltage drop across theresistor 26. In particular, for example with I₁=20 μA, i.e., I₁·R₁=200mV, the current I_(LED)=(V_(FB)−(I₁·R₁))/R_(S) circulating in the stringof LEDs 2 and in the sensing resistor 4 is zero.

The voltage drop on the resistor 4 as a result of the current signal I₁is considered negligible.

By what has been said herein, it may be noted that, in the absence ofthe current signal I₁, the current I_(LED) has a substantially constantvalue, and the load seen by an electronic transformer, in theseconditions, has a negative impedance.

Instead, in the presence of the current signal I₁, the currentcirculating in the LEDs 2 that traverses the sensing resistance 4 ismodulated in such a way that the load seen by the electronic transformerresembles a resistive load.

The present applicant has found that, to emulate a resistive load, it isexpedient for the current signal I₁ to assume values inverselyproportional to the respective values assumed by the input signalV_(IN). In other words, the current signal I₁ has a time plot 180°phase-shifted with respect to the time plot of the input voltage signalV_(IN).

The current signal I₁ that implements what has been described above isgenerated by a signal-generating circuit illustrated in FIG. 4.

In FIG. 3a , an additional biasing circuit named a current holdercircuit (CH) is shown. This circuit CH connects a resistor R_(CURR HOLD)between the terminal VIN+ and VIN− when the current I1 from currentgenerator 22 is higher than a certain value, so that the electronictransformer providing the voltage V_(IN) is loaded also when the currentrequested by the voltage regulator 5 is very low (i.e. when a rectifiedinput voltage VIN_R (see FIG. 4) is at its minimum and current I1 is atits highest value). The present applicant has found that the currentholder circuit CH further improves the emulation of a resistive load,since it loads the electronic transformer providing the voltage V_(IN)with an adequate resistor when the current signal I1 is at its maximum,i.e. when the voltage regulator 5 of the SMPS type is absorbing zerocurrent from the electronic transformer. Moreover, the current holdercircuit CH sustains the electronic transformer switching activity duringthe light load phase, so that the current generator 22 is properlybiased on the beginning of every power line cycle.

With reference to FIG. 4, the generator 22 includes a rectifier inputstage 30, for example obtained by a diode bridge 31-34, configured toreceive the input voltage V_(IN) (a.c. signal) on its own inputterminals 30 a and 30 b, and generate a rectified input voltage V_(IN)_(_) _(R) (i.e., a direct current (d.c.) signal) on its own outputterminals 30 c and 30 d.

Furthermore, the generator 22 includes a division stage 42, which iscoupled between the output terminals 30 c, 30 d of the rectifier 30 andis configured to acquire the rectified input voltage V_(IN) _(_) _(R)and generate a first intermediate operating voltage V_(P1) that is afunction of the rectified input voltage V_(IN) _(_) _(R) but has areduced maximum amplitude, in particular having a value such as to drivea first transistor 56 (operation of which is described more fullyhereinafter) into the on state. For this purpose, the division stage 42includes a resistive voltage divider formed by resistors 36, 38connected together in series between the output terminals 30 c, 30 d ofthe rectifier 30, and a capacitor 40, which is electrically coupled inparallel to the resistor 38 and has the function of providing a filterfor removal of the high frequencies (e.g., frequencies higher than60-100 kHz). The first intermediate operating voltage V_(P1), whichbiases the control terminal (gate) of the first transistor 56, is pickedup on a node 37, between the resistor 36 and the resistor 38.

By way of example, the resistor 36 has a resistance of 10 kΩ, theresistor 38 has a resistance of 2.4 kΩ, and the capacitor 40 has acapacitance of 68 nF.

The generator 22 further includes an integration stage 50, configured toreceive the first intermediate operating voltage V_(P1) and generate asecond intermediate operating voltage V_(P2) that is the integral of thefirst intermediate operating voltage V_(P1). The second intermediateoperating voltage V_(P2) is used for biasing the control terminal (base)of a second transistor 58 (operation of which is described more fullyhereinafter). For this purpose, the integration stage 50 includes: aresistor 44, electrically coupled between the node 37 and the controlterminal of the second transistor 58 (i.e., electrically coupled to theoutput terminal 30 d of the rectifier 30 via the resistor 36); and acapacitor 48 electrically coupled between the control terminal of thesecond transistor 58 and the output terminal 30 c of the rectifier 30.

By way of example, the resistor 44 has a resistance of 100 kΩ, and thecapacitor 48 has a capacitance of 1 μF.

The transistors 56 and 58 are, in particular, BJTs of a PNP type thatare the same as one another, and implement a differential pair, of a perse known type. Both the emitter terminal of the transistor 56 and theemitter terminal of the transistor 58 are electrically coupled to theoutput terminal 30 c of the rectifier 30 by a tail resistor 59, havingfor example a resistance of 43 kΩ. Furthermore, each transistor 56, 58has a respective degeneration resistor 60, 62 coupled between its ownemitter terminal and the tail resistor 59. The degeneration resistors60, 62 have the same value of resistance, for example of 30 kΩ.

The collector terminal of the transistor 56 is, for example,electrically coupled to the output terminal 30 d of the rectifier 30,whereas the collector terminal of the transistor 58 is electricallycoupled between the feedback input terminal 1 c of the SMPS device 5 andthe resistor 26 (on the node designated by the reference number 70). AZener diode (not illustrated) may likewise be coupled in parallel to theresistor 26 for providing protection from overvoltages.

FIGS. 5A-5E show, using the same time scale, voltage/current signals atinput to, and generated by, the generator 22 of FIG. 4.

FIG. 5A illustrates by way of example the envelope of the input signalV_(IN), generated by the electronic transformer, whereas FIG. 5Billustrates, by way of example, the envelope of the rectified inputsignal V_(IN) _(_) _(R) referred to the node 30 c, present on the outputof the rectifier 30.

FIG. 5C illustrates the control signals of the transistors 56 and 58referred to the node 30 c. In particular, it may be noted that the plotof the first intermediate operating voltage V_(P1) follows the plot ofthe envelope of the rectified input signal V_(IN) _(_) _(R) with a peakvalue, in modulus, lower than that of the rectified input signal V_(IN)_(_) _(R) (in this example, it ranges between 0V and −3V approximately).The second intermediate operating voltage V_(P2) is, as has been said,the integral of the first intermediate operating voltage V_(P1) and, inthis example, assumes values close to −2V.

With reference to FIG. 5C, there may be noted two operating conditionsof the differential pair. In a first operating condition, in which therectified input voltage V_(IN) _(_) _(R) has, in modulus, a maximumvalue, the differential stage does not inject current into the node 70;instead, when the rectified input voltage V_(IN) _(_) _(R) has, inmodulus, a minimum value, the differential stage injects into the node70 the current that flows through the transistor 58. According to oneembodiment, this current is the current I₁ identified previously, havinga value, in modulus, of approximately 20 μA.

FIG. 5D illustrates the plot of the currents through the transistor 56(intermediate current signal I_(INT1)) and through the transistor 58(intermediate current signal I_(INT2) corresponding to the currentsignal I₁ of FIG. 3). The sum of I_(INT1) and I_(INT2) is equal to thecurrent circulating in the resistor 59 (signal I_(INT3)). As may benoted, when the rectified input voltage V_(IN) _(_) _(R) has a maximumvalue (in modulus), the current signal I₁=I_(INT2) is minimum andapproximately to 0 A. Instead, when the rectified input voltage V_(IN)_(_) _(R) has a minimum value (in modulus), the transistor 56 is off(V_(P1)=0 V), and the transistor 58 behaves like a current generatorthat generates a current I₁ equal to approximately −20 μA, injectinginto the node 70 a current I₁ equal, in modulus, to approximately 20 μA,and thus there is a voltage drop of 200 mV on the resistor 26.

It is evident that, in the transitions of the rectified input voltageV_(IN) _(_) _(R) between the maximum value and the minimum value, thecurrent I₁ injected into the node 70 assumes intermediate values, butalways inversely proportional to the value assumed by the rectifiedinput voltage V_(IN) _(_) _(R).

FIG. 5E illustrates the voltage drop on the resistor 26, proportional tothe values assumed by the current I₁. Assuming the voltage on the node 1c set by the regulation loop of the SMPS converter as being fixed, it isevident that the current that flows in the sensing resistor 4 follows,in a directly proportional way, the variations of the input voltageV_(IN).

FIG. 4a shows a further embodiment of the present disclosure includingthe same generator 22 of FIG. 4 with the addition of a possibleimplementation of the Current holder circuit CH of FIG. 3a , which isdesignated CH4A in the embodiment of FIG. 4a . This circuit CH4Aincludes a resistor Rc1, a resistor Rc2, a resistor Rc3, a resistor Rc4,a BJT of NPN polarity Q1, a diode D1 and a MOSFET of N polarity M1, anda resistor R_(CURR) _(_) _(HOLD).

By way of example, the resistor Rc1 has a resistance of 100 kΩ, theresistor Rc2 has a resistance of 10 Ω, the resistor Rc3 has a resistanceof 10 kΩ, the resistor Rc4 has a resistance of 33 kΩ and the resistorRCURR_HOLD has a resistance of 5.1 Ω.

In particular, in the operating condition of the differential pair 56,58, when 56 is off (i.e. V_(IN) _(_) _(R) is at its minimum), the BJT Q1has no current injected in its base and therefore there is no currentflowing in the collector of Q1 and in the resistor Rc3. As aconsequence, the MOSFET M1 works with the gate equal to V_(IN) _(_) _(R)and connects with a low impedance the drain of M1 to V_(IN) _(_) _(R).The current flowing in the resistor RCURR_HOLD can be calculatedaccording to the equation

$I_{R{CURR\_ HOLD}} = \frac{V_{IN\_ R}}{R_{CURR\_ HOLD} + R_{{DSON},{M\; 1}}}$

Otherwise, in the operating condition of the differential pair 56, 58when 56 is on (i.e. V_(IN) _(_) _(R) is at its maximum), the base of Q1is biased by the current flowing in 56. The current flowing in theresistor Rc3 through (the collector of Q1) turns off M1. As a result thecurrent flowing in the resistor RCURR_HOLD is equal to zero.

The behavior of the current generator 22 and current holder circuit CH4Adescribed with reference to FIG. 4a can be seen in FIGS. 5F-5H.

When VP2 is higher than VP1 the base of transistor Q1 is positivelybiased so that there is current flowing in RC3 so that VGS of M1 goesbelow the transistor threshold voltage, thus disconnecting RCURR_HOLDfrom the electronic transformer. In this condition no resistive load isnecessary since the SMPS 5 is absorbing significant current from theelectronic transformer. Otherwise, when VP2 is lower than VP1, thecurrent IINT1 is reducing down to the condition when the base of Q1 isno more positively biased. At this point, the gate to source voltage oftransistor M1 goes above the transistor threshold voltage, thusconnecting the resistor RCURR_HOLD between the two output terminals ofthe electronic transformer. As a consequence, during this phase, asinusoidal current is absorbed from the electronic transformer.

The current holder circuit CH4A described above adds a resistive load tothe electronic transformer when the current I_(INT2) is at its maximum(i.e. when VIN_R is at its minimum). In this biasing condition, the SMPSis absorbing no current from the electronic transformer, and therefore,the connection of this resistor improves the resistive emulation of thecircuit 22. Moreover, the current holder sustains the switching activityof the electronic transformer at the beginning of every power linecycle, so that the current generator 22 works properly in every powerline cycle.

FIG. 6 shows a further embodiment of the present disclosure. Elements ofFIG. 6 common to elements appearing in FIG. 4, and described withreference to this figure, are designated by the same reference numbersand are not described any further.

According to the embodiment of FIG. 6, the generator 22 further includesa stage for biasing the tail resistor 59 of the differential stage. Forinstance, said biasing is obtained by a charge pump 75 operativelycoupled to the electronic transformer for receiving the input signalV_(IN). The charge pump 75 thus receives the input signal V_(IN) andsupplies a biasing signal V_(IN P) at input to the tail resistor 59, andis likewise electrically coupled to the node 30 c via a capacitor 83(e.g., with a capacitance of 220 nF). According to one embodiment, thetail resistor 59 of the differential stage is biased with a voltageV_(IN) _(_) _(P) having a value, in modulus, of approximately 5 V (inthis example, V_(IN) _(_) _(P)=−5 V).

The embodiment of FIG. 6 has the advantage of maintaining constant thecurrent circulating in the resistor 59 as the input signal varies andthus the linearity of the response of the current generator 22increases. As a consequence, the resistive emulation of the currentabsorbed by the SMPS 5 is improved, and the compatibility between theelectronic transformer and the SMPS is increased.

FIG. 6a illustrates a further circuit implementation of the biasing anddriving circuitry of FIG. 3a including the current generator 22 of FIG.6 and another embodiment of the current holder circuit CH of FIG. 3a ,which is designated CH6A in FIG. 6a according to yet another embodimentof the present disclosure. The structure of the current holder circuitCH6A is similar to the structure of the current holder circuit CH4A ofFIG. 4A except the resistor RC3 is coupled to the charge pump 75 toreceive the biasing signal V_(IN) _(_) _(P). The operation of thecurrent holder circuit CH6A is also similar to that of the currentholder circuit CH4A of FIG. 4A, and will be understood by those skilledin the art in view of the description of the circuit CH4A above.Briefly, when the transistor 56 of the differential pair 56, 58 isturned OFF, which occurs when the rectified input voltage V_(IN) _(_)_(R) is at its minimum, then transistor Q1 has no current injected intoits base and therefore the current through the collector of thistransistor and thus through the resistor RC3 is negligible. As a result,the transistor M1 receives approximately the voltage V_(IN) _(_) _(R) atit gate, turning ON the transistor and thereby connecting the resistorRCURR_HOLD across the rectified input voltage V_(IN) _(_) _(R) (i.e.,connecting resistor RCURR_HOLD across terminals 30 c and 30 d). Thecurrent I_(RCURR) _(_) _(HOLD) through the resistor RCURR_HOLD is againgiven by the above equation. Conversely, when the rectified inputvoltage V_(IN) _(_) _(R) has its maximum value, current from transistor56 turns ON the transistor Q1 which, in turn, drives the voltage appliedto the gate of the transistor M1 to a voltage level that turns thetransistor M1 OFF. In this situation no meaningful current flows throughthe resistor RCURR_HOLD as this resistor is effectively isolated fromthe rectified input voltage V_(IN) _(_) _(R) by the deactivatedtransistor M1.

FIG. 6b illustrates a still further circuit implementation of thebiasing and driving circuit 20 of FIG. 3a according to yet anotherembodiment of the present disclosure. In this embodiment, the pumpedvoltage VIN_P generated by the charge pump 75 is supplied to bias theonly current holder circuit CH6A. This is in contrast to the embodimentof FIG. 6 where the pumped voltage VIN_P is applied to bias only thecurrent generator 22 and the embodiment of FIG. 6a where the pumpedvoltage VIN_P is applied to bias both the current generator 22 and thecurrent holder circuit CH6A. The use of the pumped voltage VIN_P has thebenefit of increasing the voltage biasing of M1 gate, so it helpsconnecting the resistive load at the output of the electronictransformer when the voltage VIN_R is at its very minimum, i.e. at thebeginning of every power line cycle.

FIG. 7 shows a circuit embodiment of the charge pump 75 of FIG. 6.Elements of the circuit of FIG. 7 that are in common with those of thecircuit of FIG. 6 are designated by the same reference numbers and arenot described any further. The charge pump 75 includes a diode 76 and aresistor 78 (e.g., with a resistance of 1 kΩ), connected together inseries between the input terminal at the voltage V_(IN) ⁻ (groundreference GND) and an intermediate node 79; in particular, the diode 76has its anode coupled to V_(IN) ⁻ and its cathode coupled to theresistor 78. Furthermore, the charge pump 75 includes a capacitor 80(e.g., with a capacitance of 220 nF) and a Zener diode 81 coupled inparallel to one another, between the intermediate node 79 and the inputterminal at the voltage V_(IN) ⁺; in particular, the Zener diode 81 hasits anode coupled to V_(IN) ⁺ and its cathode coupled to theintermediate node 79. A diode 82, having its anode coupled to theintermediate node 79, is set on the output of the charge pump 75, forsupplying at output the signal V_(IN) _(_) _(P).

The advantages obtained emerge clearly from the foregoing description.

In particular, the biasing and driving circuit described may be used forany generic SMPS, enabling operative coupling of said generic SMPS witha generic electronic transformer that requires a resistive load at theoutput of the transformer. Consequently, the power factor is increased.

The biasing and driving circuit described further supports SMPSs withboth current-mode and voltage-mode internal architecture.

Modifications and variations may be made to the device and to the methoddescribed herein, without thereby departing from the scope of thepresent disclosure, as defined in the annexed claims.

In particular, the present disclosure applies to any generic feedbackvoltage regulator (whether of the SMPS switching type or of the lineartype).

Furthermore, the driven electric load may be a generic electric load,not limited to the string of LEDs.

The various embodiments described above can be combined to providefurther embodiments. All of the U.S. patents, U.S. patent applicationpublications, U.S. patent applications, foreign patents, foreign patentapplications and non-patent publications referred to in thisspecification and/or listed in the Application Data Sheet areincorporated herein by reference, in their entirety. Aspects of theembodiments can be modified, if necessary to employ concepts of thevarious patents, applications and publications to provide yet furtherembodiments.

These and other changes can be made to the embodiments in light of theabove-detailed description. In general, in the following claims, theterms used should not be construed to limit the claims to the specificembodiments disclosed in the specification and the claims, but should beconstrued to include all possible embodiments along with the full scopeof equivalents to which such claims are entitled. Accordingly, theclaims are not limited by the disclosure.

The invention claimed is:
 1. A biasing and driving circuit for anelectric load, having input terminals configured to receive an a.c.input voltage and output terminals configured to supply a d.c. outputvoltage to said electric load, comprising: a voltage regulator with afeedback loop having a feedback input configured to receive a sensingvoltage that is a function of a supply current that flows through theelectric load and for regulating, on the basis of the sensing voltagereceived, said supply current; and a resistive sensing element,operatively coupled to the feedback input, configured to receive thesupply current and generate said sensing voltage as a function of saidsupply current received; a current transducer, of a resistive type,coupled to the feedback input; and an auxiliary biasing circuitconfigured to receive said a.c. input voltage and to inject through saidcurrent transducer an a.c. auxiliary biasing current that varies in away inversely proportional to the a.c. input voltage.
 2. The biasing anddriving circuit according to claim 1, wherein the auxiliary biasingcircuit includes: a rectifier, configured to receive the a.c. inputvoltage, rectify the a.c. input voltage, and to supply at an output arectified voltage; a differential stage, having a first output coupledto the feedback input for supplying the a.c. auxiliary biasing current;a voltage-divider unit, configured to receive the rectified voltage andsupply a first operating voltage, obtained as a partition of therectified voltage, to a first input of the differential stage; and anintegrator unit, configured to receive the divided voltage and supply toa second input of the differential stage a second operating voltageobtained as integral of the first operating voltage.
 3. The biasing anddriving circuit according to claim 2, wherein the differential stageincludes a first transistor and a second transistor, driven respectivelyby the first and second operating voltages, outputting the a.c.auxiliary biasing current which has a first operating value when therectified voltage has a value maximum in modulus; and a second operatingvalue, higher than the first operating value, when the rectified voltagehas a value minimum in modulus.
 4. The biasing and driving circuitaccording to claim 3, wherein a bandwidth of the feedback loop is higherthan a maximum frequency of the a.c. auxiliary biasing current.
 5. Thebiasing and driving circuit according to claim 4, wherein the voltageregulator is a regulator of an SMPS type, said feedback loop comprisingan error amplifier coupled to the feedback input, having a first inputconfigured to receive the sensing voltage, a second input configured toreceive a feedback reference voltage, and an output for supplying acontrol signal, said first operating value being a current value equalto substantially zero, and the second operating value being a currentvalue such that said current transducer biases the feedback input at avoltage value equal to the feedback voltage reference of the erroramplifier.
 6. The biasing and driving circuit according to claim 2,further comprising a charge pump configured to receive the a.c. inputvoltage and to generate a biasing voltage configured to bias saiddifferential stage.
 7. The biasing and driving circuit according toclaim 6, further comprising: a first controllable switching elementcoupled in series with a first resistive element across the output ofthe rectifier; a second controllable switching element coupled in serieswith a second resistive element between one side of the output of therectifier and the charge pump to receive the biasing voltage, a nodebeing defined at the interconnection of the second controllableswitching element and the second resistive element and the node beingcoupled to the first controllable switching element; and wherein thedifferential stage includes a second output that controls the secondcontrollable switching element to selectively activate the firstcontrollable switching element to couple the first resistive elementacross the output of the rectifier or to isolate the resistive elementfrom the output of the rectifier.
 8. The biasing and driving circuitaccording to claim 2, further comprising a current holder circuitcomprising: a controllable switching element coupled in series with aresistive element across the output of the rectifier; and wherein thedifferential stage includes a second output that controls thecontrollable switching element to couple the resistive element acrossthe output of the rectifier or to isolate the resistive element from theoutput of the rectifier.
 9. The biasing and driving circuit according toclaim 2, further comprising: a charge pump configured to receive thea.c. input voltage and to generate a biasing voltage; a firstcontrollable switching element coupled in series with a first resistiveelement across the output of the rectifier and coupled to the chargepump to receive the biasing voltage to bias the first controllableswitching element; and a second controllable switching element coupledin series with a second resistive element between one side of the outputof the rectifier and the charge pump to receive the biasing voltage, anode being defined at the interconnection of the second controllableswitching element and the second resistive element and the node beingcoupled to the first controllable switching element.
 10. The biasing anddriving circuit according to claim 1, wherein said electric loadcomprises a string of light-emitting devices.
 11. The biasing anddriving circuit according to claim 1, further comprising a currentholder circuit configured to couple a resistive element between theinput terminals responsive to the a.c. auxiliary biasing current beinggreater than a threshold value.
 12. The biasing and driving circuitaccording to claim 1, wherein the a.c. input voltage is a sinusoidalsignal and the auxiliary biasing current is a sinusoidal signal havingan approximately 180 degree phase shift relative to the a.c. inputvoltage.
 13. An electronic system, comprising: an electronic load; afeedback voltage regulator circuit having a feedback loop including afeedback input terminal configured to receive a sensing voltage that isa function of a supply current through the electronic load, and thefeedback voltage regulator configured to regulate, on the basis of thesensing voltage, the supply current; a resistive sensing element,operatively coupled to the feedback input terminal, configured toreceive the supply current and to generate the sensing voltage as afunction of the supply current; a current transducer coupled to thefeedback input terminal; and an auxiliary biasing circuit configured toreceive an a.c. input voltage and to supply through the currenttransducer an a.c. auxiliary biasing current that is inverselyproportional to the a.c. input voltage.
 14. The electronic system ofclaim 13, wherein the feedback voltage regulator circuit comprises oneof a switching-mode power supply (SMPS) and a linear power supply. 15.The electronic system of claim 13, wherein the electronic load comprisesat least one light-emitting diode.
 16. The electronic system of claim13, wherein the auxiliary biasing circuit is external to an integratedcircuit that includes the feedback voltage regulator circuit.
 17. Amethod of controlling a feedback voltage regulator circuit, the methodcomprising: generating a sensing voltage as a function of a supplycurrent through an electronic load coupled to the feedback voltageregulator circuit; coupling the sensing voltage to a feedback input ofthe feedback voltage regulator circuit to generate a feedback voltage onthe feedback input; generating an a.c. auxiliary biasing current that isinversely proportional to an a.c. input voltage; and providing the a.c.auxiliary biasing current to the feedback input to thereby control thevalue of the feedback voltage on the feedback input.
 18. The method ofclaim 17, further comprising: rectifying the a.c. input voltage; andsupplying the rectified a.c. input voltage to a supply input of thefeedback voltage regulator circuit.
 19. The method of claim 18, furthercomprising utilizing a differential pair of transistors to generate thea.c. auxiliary biasing current that is provided to the feedback input.20. The method of claim 19, further comprising: generating a pumpedvoltage from the a.c. input voltage; and providing the pumped voltage tobias the differential pair of transistors.
 21. The method of claim 20further comprising coupling a resistive element across to the rectifieda.c. input voltage responsive to the a.c. auxiliary biasing currentbeing greater than a threshold value.